There is a continuing industry demand for converters of increasing power density, which means more power transferred in a given volume. A method for increasing the power transfer through a converter is to increase the switching frequency in order to minimize the size of the transformer and the capacitors. Using prior art topologies such as forward or flyback, which employ “hard” switching techniques, makes high frequency operation less efficient. The switching losses associated with switching elements that turn on when there is a voltage across them are proportional to the switching frequency. An increase in switching frequency leads to an increase in switching losses and an increase in the level of electromagnetic interference (EMI).
In order to overcome limitations in switching speeds, the prior art has devised a new family of soft transition. The U.S. Pat. Nos. 5,132,889, 5,126,931, 5,231,563 and 5,434,768 present several methods of accomplishing zero voltage across the primary switches at switching. These patents are incorporated herein by reference.
Additional power loss in converters is due to the reverse recovery in output rectifiers. During switching when a negative polarity voltage is applied to a rectifier that is in conduction, the current through the rectifier will continue to conduct until all the carriers in the rectifier's junctions are depleted. During this period of time, the current polarity will reverse the current flowing from the cathode to the anode, while the voltage across the diode is still positive from the anode to the cathode. The current flowing in reverse through the diode will reach a peak value referred to in literature as Irrm. Further on, while the rectifiers' junction is depleting the carriers, the rectifier becomes a high impedance device. The current through the rectifier will decrease rapidly from Irrm level to zero. During the same time, the negative voltage across the rectifier will build up to high levels.
During the period of time when there is a negative voltage across the diode and negative current is flowing through it, there will be power dissipation in the device. This kind of loss is referred in the literature as reverse recovery loss. The reverse recovery loss is proportional with the reverse recovery current Irrm, the negative voltage across the rectifier and the frequency.
The reverse recovery current Irrm, which is a key component in reverse recovery loss, is a function of the type of device, the temperature and the current slope at turn off. The reverse recovery characteristics are worse for higher voltage rectifiers. As a result, the reverse recovery loss becomes a significant loss mechanism for higher output voltage applications. The reverse recovery current Irrm is directly dependent on the current slope at turn off. A “soft” or low current slope reduces the reverse recovery current and as a consequence reduces the reverse recovery loss. To accomplish a very soft slope current at turn off, an inductive element has to be in series with the rectifier. The inductor element will prevent a fast current variation dI/dt. However, the presence of an inductive element in series with the rectifier will increase the negative voltage across the rectifier at turn off. The reverse voltage across the rectifier can reach very high levels and can exceed the voltage break down of the device, leading to failure.
RC snubbers or complicated lossless snubbers can be added across the rectifier to reduce the reverse recovery loss and the voltage stress on the devices. This leads to complex circuits and this negatively affects efficiency and reliability. As a result of these limitations, the high voltage converters have had to operate at lower frequency in order to reduce the power dissipation associated with reverse recovery.
FIG. 2A illustrates a standard full bridge phase shifted topology. The primary switching elements, M1, M2, M3 and M4 are controlled as depicted in the key waveforms of FIG. 2B. During the time M1 and M4 are conducting, there is a positive voltage at the dot A in the secondary winding and the rectifier D1 is conducting. When M4 turns off, in the primary winding the current will continue to conduct, and its path will be through the parasitic capacitance of M3 and M4, discharging the parasitic capacitance across M3 to zero and as a result creating zero voltage switching conditions. As a result, M3 will turn on at zero voltage. Further, M1 and M3 will be conducting. During this time, the primary winding of the transformer is shorted and the voltage in the secondary winding is zero. Both D1 and D2 will conduct during this time. The current through Lo will be split equally between D1 and D2.
At the moment when M1 turns off, the current will continue to flow in the primary discharging the parasitic capacitance of M2 towards zero. If certain conditions are met, M2 will turn on at zero voltage conditions. In any event, when M3 and M2 conduct, the polarity will change in the secondary winding. The change in polarity will force the current flowing through Lo to flow totally through D2 and the rectifier D1 will be reverse biased. Due to the reverse recovery characteristic of D1, the current will flow in reverse through D1 as shown at point 5 until the carriers in the junction are depleted. After that the rectifier D1 will behave as a high impedance device. As a function of the current slope through D1 at turn off, which determines the reverse recovery characteristics of D1, and as a function of the parasitic inductive elements Lo in series with D1, large voltage spikes will develop across D1 as shown at point 10 in FIG. 2B. This phenomenon will lead to reverse recovery losses in D1, when the reverse recovery current and reverse voltage will be present on the rectifier. In addition, these large voltage spikes developed across D1 may lead to voltage stress, which may exceed the rating of the device. For this reason, snubbing circuits may have to be employed across D1, which will increase the power dissipation, increase circuit complexity, decrease reliability and decrease power density. The reverse recovery current associated with D1 will also create a “temporary short” across the secondary winding, preventing the resonant transition across M2 from achieving zero voltage switching conditions. The voltage ringing across D1 will also lead to increased EMI.
The losses associated with the reverse recovery of the output rectifiers are proportionate with the frequency. The trend towards miniaturization requires an increase in switching frequency, which leads to more reverse recovery losses. In addition, in order to accomplish minaturization, higher efficiency in converters is necessary to minimize heat. In conclusion then, there remains a need for a topology wherein the reverse recovery losses are eliminated, in this way allowing an increase in switching frequency without a penalty in efficiency.